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LT3757 Boost, Flyback, SEPIC and Inverting Controller DESCRIPTION
The LT(R)3757 is a wide input range, current mode, DC/DC controller which is capable of generating either positive or negative output voltages. It can be configured as either a boost, flyback, SEPIC or inverting converter. The LT3757 drives a low side external N-channel power MOSFET from an internal regulated 7.2V supply. The fixed frequency, current-mode architecture results in stable operation over a wide range of supply and output voltages. The operating frequency of LT3757 can be set with an external resistor over a 100kHz to 1MHz range, and can be synchronized to an external clock using the SYNC pin. A low minimum operating supply voltage of 2.9V, and a low shutdown quiescent current of less than 1A, make the LT3757 ideally suited for battery-operated systems. The LT3757 features soft-start and frequency foldback functions to limit inductor current during start-up and output short-circuit. The device is available in a small 10-lead DFN (3mm x 3mm) or MSOP package.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.
FEATURES
n n n n n n n n n n n
Wide Input Voltage Range: 2.9V to 40V Positive or Negative Output Voltage Programming with a Single Feedback Pin Current Mode Control Provides Excellent Transient Response Programmable Operating Frequency (100kHz-1MHz) with One External Resistor Synchronizeable to an External Clock Output Overvoltage Protection Low Shutdown Current < 1A Internal 7.2V Low Dropout Voltage Regulator Programmable Input Undervoltage Lockout with Hysteresis Programmable Soft-Start Small 10-Lead DFN (3mm x 3mm) and Thermally Enhanced 10-Pin MSOP Packages
APPLICATIONS
n n n
Automotive and Industrial Boost, Flyback, SEPIC and Inverting Converters Telecom Power Supplies Portable Electronic Equipment
TYPICAL APPLICATION
High Efficiency Boost Converter
VIN 8V TO 16V 10F 25V X5R 100 200k VIN SHDN/UVLO 43.2k SYNC RT SS VC 41.2k 300kHz 0.1F 6.8nF 22k 10H VOUT 24V 2A GATE SENSE FBX GND INTVCC 4.7F 10V X5R 226k 90 80 EFFICIENCY (%) 70 60 50 10F 25V X5R 40 30 0.001 VIN = 8V VIN = 16V
Efficiency
LT3757
+
16.2k 0.01
47F 35V x2
0.1 1 0.01 OUTPUT CURRENT (A)
10
3757 TA01b
3757 TA01a
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ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN, SHDN/UVLO .......................................................40V INTVCC .................................................... VIN + 0.3V, 20V GATE .........................................................INTVCC + 0.3V SYNC ..........................................................................8V VC, SS.........................................................................3V RT ............................................................................1.5V
SENSE....................................................................0.3V FBX ................................................................. -6V to 6V Operating Temperature Range (Note 2) ............................................. -40C to 125C Maximum Junction Temperature........................... 125C Storage Temperature Range................... -65C to 125C
PIN CONFIGURATION
TOP VIEW TOP VIEW VC FBX SS RT SYNC 1 2 3 4 5 11 10 VIN 9 SHDN/UVLO 8 INTVCC 7 GATE 6 SENSE VC FBX SS RT SYNC 1 2 3 4 5 10 9 8 7 6 VIN SHDN/UVLO INTVCC GATE SENSE
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MSE PACKAGE 10-LEAD PLASTIC MSOP TJMAX = 125C, JA = 40C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
DD PACKAGE 10-LEAD (3mm 3mm) PLASTIC DFN TJMAX = 125C, JA = 43C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LT3757EDD#PBF LT3757IDD#PBF LT3757EMSE#PBF LT3757IMSE #PBF TAPE AND REEL LT3757EDD#TRPBF LT3757IDD#TRPBF LT3757EMSE#TRPBF LT3757IMSE#TRPBF PART MARKING* LDYW LDYW LTDYX LTDYX PACKAGE DESCRIPTION 10-Lead (3mm x 3mm) Plastic DFN 10-Lead (3mm x 3mm) Plastic DFN 10-Lead (3mm x 3mm) Plastic MSOP 10-Lead (3mm x 3mm) Plastic MSOP TEMPERATURE RANGE -40C to 125C -40C to 125C -40C to 125C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
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ELECTRICAL CHARACTERISTICS
PARAMETER VIN Operating Range VIN Shutdown IQ VIN Operating IQ VIN Operating IQ with Internal LDO Disabled SENSE Current Limit Threshold SENSE Input Bias Current Error Amplifier FBX Regulation Voltage (VFBX(REG)) FBX Overvoltage Lockout FBX Pin Input Current Transconductance gm (IVC /VFBX) VC Output Impedance VFBX Line Regulation (VFBX /VIN ) VC Current Mode Gain (VVC /VSENSE) VC Source Current VC Sink Current Oscillator Switching Frequency
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 24V, SHDN/UVLO = 24V, FBX = 0V, SENSE = 0V, unless otherwise noted.
CONDITIONS SHDN/UVLO = 0V SHDN/UVLO = 1.15V VC = 0.3V, RT = 41.2k VC = 0.3V, RT = 41.2k, INTVCC = 7.5V
l
MIN 2.9
TYP 0.1 1.6 280
MAX 40 1 6 2.2 400 120
UNITS V A A mA A mV A
100
110 -65
Current Out of Pin VFBX > 0V (Note 3) VFBX < 0V (Note 3) VFBX > 0V (Note 4) VFBX < 0V (Note 4) VFBX = 1.6V (Note 3) VFBX = -0.8V (Note 3) (Note 3) (Note 3) VFBX > 0V, 2.9V < VIN < 40V (Note 3) VFBX < 0V, 2.9V < VIN < 40V (Note 3) VFBX = 0V, VC = 1.5V VFBX = 1.7V VFBX = -0.85V RT = 41.2k to GND, VFBX = 1.6V RT = 140k to GND, VFBX = 1.6V RT = 10.5k to GND, VFBX = 1.6V VFBX = 1.6V 270
l l
1.569 -0.816 6 7 -10
1.6 -0.80 8 11 70 230 5 0.003 0.002 5.5 -15 12 11 300 100 1000 1.2 220 220
1.631 -0.784 10 14 100 10
V V % % nA nA S M
0.09 0.04
% % V/V A A A
330
kHz kHz kHz V ns ns
RT Voltage Minimum Off-Time Minimum On-Time SYNC Input Low SYNC Input High SS Pull-Up Current Low Dropout Regulator INTVCC Regulation Voltage INTVCC Undervoltage Lockout Threshold INTVCC Overvoltage Lockout Threshold INTVCC Current Limit INTVCC Load Regulation (VINTVCC / VINTVCC) INTVCC Line Regulation VINTVCC /(VINTVCC * VIN) Dropout Voltage (VIN - VINTVCC)
0.4 1.5 SS = 0V, Current Out of Pin
l
V V A
-10 7 2.6 7.2 2.7 0.1 17.5 40 95 -0.5 0.008 400 0.03 7.4 2.8
V V V V mA mA % %/V mV
Falling INTVCC UVLO Hysteresis VIN = 40V VIN = 15V 0 < IINTVCC < 20mA, VIN = 8V 8V < VIN < 40V VIN = 6V, IINTVCC = 20mA
30 -0.9
55
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ELECTRICAL CHARACTERISTICS
PARAMETER INTVCC Current in Shutdown INTVCC Voltage to Bypass Internal LDO Logic Inputs SHDN/UVLO Threshold Voltage Falling SHDN/UVLO Input Low Voltage SHDN/UVLO Pin Bias Current Low SHDN/UVLO Pin Bias Current High Gate Driver t r Gate Driver Output Rise Time t f Gate Driver Output Fall Time Gate VOL Gate VOH
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 24V, SHDN/UVLO = 24V, FBX = 0V, SENSE = 0V, unless otherwise noted.
CONDITIONS SHDN/UVLO = 0V, INTVCC = 8V MIN TYP 16 7.5 VIN = INTVCC = 8V I(VIN) Drops Below 1A SHDN/UVLO = 1.15V SHDN/UVLO = 1.30V CL = 3300pF (Note 5), INTVCC = 7.5V CL = 3300pF (Note 5), INTVCC = 7.5V INTVCC - 50mV 1.7 2 10 22 20 0.05
l
MAX
UNITS A V V V A nA ns ns V V
1.17
1.22
1.27 0.4 2.5 100
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3757E is guaranteed to meet performance specifications from the 0C to 125C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The
LT3757I is guaranteed over the full -40C to 125C operating junction temperature range. Note 3: The LT3757 is tested in a feedback loop which servos VFBX to the reference voltages (1.6V and -0.8V) with the VC pin forced to 1.3V. Note 4: FBX overvoltage lockout is measured at VFBX(OVERVOLTAGE) relative to regulated VFBX(REG). Note 5: Rise and fall times are measured at 10% and 90% levels.
TYPICAL PERFORMANCE CHARACTERISTICS
Positive Feedback Voltage vs Temperature, VIN
1602 REGULATED FEEDBACK VOLTAGE (mV) 1600 VIN = 24V 1598 1596 VIN = 8V 1594 1592 1590 1588 -50 VIN = INTVCC = 2.9V REGULATED FEEDBACK VOLTAGE (mV) VIN = 40V -792 -794 -796 -798 -800 -802 -804 -50 VIN = 24V VIN = INTVCC = 2.9V
TA = 25C, unless otherwise noted.
Negative Feedback Voltage vs Temperature, VIN
1.8
Quiescent Current vs Temperature, VIN
QUIESCENT CURRENT (mA)
1.7 VIN = 40V VIN = 24V
VIN = 8V
1.6
VIN = 40V
1.5 VIN = INTVCC = 2.9V 1.4 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
-25
50 25 0 75 TEMPERATURE (C)
100
125
-25
50 25 0 75 TEMPERATURE (C)
100
125
3757 G01
3757 G02
3757 G02
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TA = 25C, unless otherwise noted. Normalized Switching Frequency vs FBX
120 NORMALIZED FREQUENCY (%) 100 80 60 40 20 0 -0.8
TYPICAL PERFORMANCE CHARACTERISTICS
Dynamic Quiescent Current vs Frequency
35 CL = 3300pF 30 25 RT (k) IQ(mA) 20 15 10 5 0 10 1000
RT vs Switching Frequency
100
0 100 200 300 400 500 600 700 800 900 1000 SWITCHING FREQUENCY (KHz)
3757 G04
0 100 200 300 400 500 600 700 800 900 1000 SWITCHING FREQUENCY (KHz)
3757 G05
-0.4
0 0.4 0.8 FBX VOLTAGE (V)
1.2
1.6
3757 G06
Switching Frequency vs Temperature
325 320 SWITCHING FREQUENCY (kHz) 315 310 305 300 295 290 285 280 275 -50 -25 0 25 50 75 TEMPERATURE (C) 100 125 RT = 41.2K 120
SENSE Current Limit Threshold vs Temperature
115
SENSE Current Limit Threshold vs Duty Cycle
SENSE THRESHOLD (mV)
SENSE THRESHOLD (mV) 50 25 0 75 TEMPERATURE (C) 125
115
110
110
105
105
100
100
95 -50
95 -25 100 0 20 40 60 DUTY CYCLE (%) 80 100
3757 G09
3757 G07
3757 G08
SHDN/UVLO Threshold vs Temperature
1.24 40
SHDN/UVLO Current vs Voltage
2.4
SHDN/UVLO Hysteresis Current vs Temperature
SHDN/UVLO CURRENT (A)
SHDN/UVLO VOLTAGE (V)
1.23
30 ISHDN / UVLO (A)
2.2
1.22
20
2.0
1.21
10
1.8
1.2 -50
0 -25 50 25 0 75 TEMPERATURE (C) 100 125 0 10 20 30 SHDN/UVLO VOLTAGE (V) 40
3757 G11
1.6 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
3757 G10
3757 G12
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TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC vs Temperature
7.4 90 80
TA = 25C, unless otherwise noted.
INTVCC Minimum Output Current vs VIN
TJ = 125C 7.3
INTVCC Load Regulation
VIN = 8V 7.2 INTVCC VOLTAGE (V)
7.3 INTVCC (V)
INTVCC CURRENT (mA)
70 60 50 40 30 20 10 INTVCC = 4.5V INTVCC = 6V
7.1
7.2
7
7.1
6.9
7.0 -50
0 -25 50 25 0 75 TEMPERATURE (C) 100 125
0
5
10
15
20 VIN (V)
25
30
35
40
6.8 0 10
40 50 30 20 INTVCC LOAD (mA)
60
70
3757 G13
3757 G14
3757 G15
INTVCC Line Regulation
7.30 700 600 INTVCC VOLTAGE (V) 7.25 DROPOUT VOLTAGE (mV) 500 400 300 200 100 7.10 0 5 10 15 20 25 VIN (V) 30 35 40
3757 G16
INTVCC Dropout Voltage vs Current, Temperature
90 125C 80 70 75C 60 25C 0C -50C TIME (ns)
Gate Drive Rise and Fall Time vs CL
INTVCC = 7.2V
RISE TIME 50 40 30 20 10 0
FALL TIME
7.20
7.15
0 0 5 10 INTVCC LOAD (mA)
3757 G17
15
20
0
5
10
15 CL (nF)
20
25
30
3757 G18
Gate Drive Rise and Fall Time vs INTVCC
30 25 20 TIME (ns) FALL TIME 15 10 5 0 3 6 9 INTVCC (V)
3757 G19
Typical Start-Up Waveforms
VIN = 12V VOUT 10V/DIV
FBX Frequency Foldback Waveforms During Overcurrent
VIN = 12V
CL = 3300pF RISE TIME VOUT 5V/DIV
VSW 20V/DIV
IL1A + IL1B 5A/DIV 2ms/DIV PAGE 31 CIRCUIT 12 15
3757 G20
IL1A + IL1B 5A/DIV
50s/DIV PAGE 31 CIRCUIT
3757 G21
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GATE (Pin 7): N-Channel MOSFET Gate Driver Output. Switches between INTVCC and GND. Driven to GND when IC is shut down, during thermal lockout or when INTVCC is above or below the OV or UV thresholds, respectively. INTVCC (Pin 8): Regulated Supply for Internal Loads and Gate Driver. Supplied from VIN and regulated to 7.2V (typical). INTVCC must be bypassed with a minimum of 4.7F capacitor placed close to pin. INTVCC can be connected directly to VIN, if VIN is less than 17.5V. INTVCC can also be connected to a power supply whose voltage is higher than 7.5V, and lower than VIN, provided that supply does not exceed 17.5V. SHDN/UVLO (Pin 9): Shutdown and Undervoltage Detect Pin. An accurate 1.22V (nominal) falling threshold with externally programmable hysteresis detects when power is okay to enable switching. Rising hysteresis is generated by the external resistor divider and an accurate internal 2A pull-down current. An undervoltage condition resets sort-start. Tie to 0.4V, or less, to disable the device and reduce VIN quiescent current below 1A. VIN (Pin 10): Input Supply Pin. Must be locally bypassed with a 0.22F or larger, capacitor placed close to the , pin. Exposed Pad (Pin 11): Ground. This pin also serves as the negative terminal of the current sense resistor. The Exposed Pad must be soldered directly to the local ground plane.
PIN FUNCTIONS
VC (Pin 1): Error Amplifier Compensation Pin. Used to stabilize the voltage loop with an external RC network. FBX (Pin 2): Positive and Negative Feedback Pin. Receives the feedback voltage from the external resistor divider across the output. Also modulates the frequency during start-up and fault conditions when FBX is close to GND. SS (Pin 3): Soft-Start Pin. This pin modulates compensation pin voltage (VC) clamp. The soft-start interval is set with an external capacitor. The pin has a 10A (typical) pull-up current source to an internal 2.5V rail. The softstart pin is reset to GND by an undervoltage condition at SHDN/UVLO, an INTVCC undervoltage or overvoltage condition or an internal thermal lockout. RT (Pin 4): Switching Frequency Adjustment Pin. Set the frequency using a resistor to GND. Do not leave this pin open. SYNC (Pin 5): Frequency Synchronization Pin. Used to synchronize the switching frequency to an outside clock. If this feature is used, an RT resistor should be chosen to program a switching frequency 20% slower than the SYNC pulse frequency. Tie the SYNC pin to GND if this feature is not used. SYNC is ignored when FBX is close to GND. SENSE (Pin 6): The Current Sense Input for the Control Loop. Kelvin connect this pin to the positive terminal of the switch current sense resistor in the source of the N-FET. The negative terminal of the current sense resistor should be connected to GND plane close to the IC.
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BLOCK DIAGRAM
L1 VIN R4 R3 CDC D1 VOUT L2 COUT2 COUT1 R1
*
+
CIN
*
9 SHDN/UVLO A10 2.5V IS3 VC 1 CC2 CC1 RC 1.72V IS1 2A 2.5V IS2 10A Q3 G4 A11 G3 INTERNAL REGULATOR AND UVLO UVLO 10 VIN
FBX
R2 +
+ -
1.22V
+ A9 -
A8
17.5V
CURRENT LIMIT 7.2V LDO INTVCC 8 CVCC
- + - +
TSD 165C G6 VC
+ -
2.7V UP 2.6V DOWN DRIVER SR1 GATE O S G2 7 M1
-0.88V
Q2
PWM COMPARATOR A6 SLOPE RAMP RAMP GENERATOR VISENSE
1.6V FBX FBX 2
+ A1 - + A2 -
-0.8V
FREQUENCY FOLDBACK
1.25V FREQ FOLDBACK
SS 3 CSS 5
Figure 1. LT3757 Block Diagram Working as a SEPIC Converter
8
+ -
1.25V
A3
SYNC 4
+ -
A12
A7
G5
R
- +
108mV
SENSE
+ A5 -
6 GND 11 RSENSE
G1
100kHz-1MHz OSCILLATOR
+ + -
A4 FREQ PROG
Q1
RT
3757 F01
RT
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is pulled down to -0.8V by a voltage divider connected from VOUT to GND. Comparator A1 becomes inactive and comparator A2 performs the noninverting amplification from FBX to VC. The LT3757 has overvoltage protection functions to protect the converter from excessive output voltage overshoot during start-up or recovery from a short-circuit condition. An overvoltage comparator A11 (with 20mV hysteresis) senses when the FBX pin voltage exceeds the positive regulated voltage (1.6V) by 8% and provides a reset pulse. Similarly, an overvoltage comparator A12 (with 10mV hysteresis) senses when the FBX pin voltage exceeds the negative regulated voltage (-0.8V) by 11% and provides a reset pulse. Both reset pulses are sent to the main RS latch (SR1) through G6 and G5. The power MOSFET switch M1 is actively held off for the duration of an output overvoltage condition. Programming Turn-On and Turn-Off Thresholds with the SHDN/UVLO Pin The SHDN/UVLO pin controls whether the LT3757 is enabled or is in shutdown state. A micropower 1.22V reference, a comparator A10 and a controllable current source IS1 allow the user to accurately program the supply voltage at which the IC turns on and off. The falling value can be accurately set by the resistor dividers R3 and R4. When SHDN/UVLO is above 0.7V, and below the 1.22V threshold, the small pull-down current source IS1 (typical 2A) is active. The purpose of this current is to allow the user to program the rising hysteresis. The Block Diagram of the comparator and the external resistors is shown in Figure 1. The falling threshold voltage and rising threshold voltage can be calculated by the following equations: (R3 + R4) R4 VVIN, RISING = 2A * R3+ VIN, FALLING VVIN, FALLING = 1.22 *
APPLICATIONS INFORMATION
Main Control Loop The LT3757 uses a fixed frequency, current mode control scheme to provide excellent line and load regulation. Operation can be best understood by referring to the Block Diagram in Figure 1. The start of each oscillator cycle sets the SR latch (SR1) and turns on the external power MOSFET switch M1 through driver G2. The switch current flows through the external current sensing resistor RSENSE and generates a voltage proportional to the switch current. This current sense voltage VISENSE (amplified by A5) is added to a stabilizing slope compensation ramp and the resulting sum (SLOPE) is fed into the positive terminal of the PWM comparator A7. When SLOPE exceeds the level at the negative input of A7 (VC pin), SR1 is reset, turning off the power switch. The level at the negative input of A7 is set by the error amplifier A1 (or A2) and is an amplified version of the difference between the feedback voltage (FBX pin) and the reference voltage (1.6V or -0.8V, depending on the configuration). In this manner, the error amplifier sets the correct peak switch current level to keep the output in regulation. The LT3757 has a switch current limit function. The current sense voltage is input to the current limit comparator A6. If the SENSE pin voltage is higher than the sense current limit threshold VSENSE(MAX) (108mV, typical), A6 will reset SR1 and turn off M1 immediately. The LT3757 is capable of generating either positive or negative output voltage with a single FBX pin. It can be configured as a boost, flyback or SEPIC converter to generate positive output voltage, or as an inverting converter to generate negative output voltage. When configured as a SEPIC converter, as shown in Figure 1, the FBX pin is pulled up to the internal bias voltage of 1.6V by a voltage divider (R1 and R2) connected from VOUT to GND. Comparator A2 becomes inactive and comparator A1 performs the inverting amplification from FBX to VC. When the LT3757 is in an inverting configuration, the FBX pin
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APPLICATIONS INFORMATION
For applications where the SHDN/UVLO pin is only used as a logic input, the SHDN/UVLO pin can be connected directly to the input voltage VIN for always-on operation. INTVCC Regulator Bypassing and Operation An internal, low dropout (LDO) voltage regulator produces the 7.2V INTVCC supply which powers the gate driver, as shown in Figure 1. If a low input voltage operation is expected (e.g., supplying power from a lithium-ion battery or a 3.3V logic supply), low threshold MOSFETs should be used. The LT3757 contains an undervoltage lockout comparator A8 and an overvoltage lockout comparator A9 for the INTVCC supply. The INTVCC undervoltage (UV) threshold is 2.7V (typical), with 100mV hysteresis, to ensure that the MOSFETs have sufficient gate drive voltage before turning on. The logic circuitry within the LT3757 is also powered from the internal INTVCC supply. The INTVCC overvoltage (OV) threshold is set to be 17.5V (typical) to protect the gate of the power MOSFET. When INTVCC is below the UV threshold, or above the OV threshold, the GATE pin will be forced to GND and the soft-start operation will be triggered. The INTVCC regulator must be bypassed to ground immediately adjacent to the IC pins with a minimum of 4.7F ceramic capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate driver. In an actual application, most of the IC supply current is used to drive the gate capacitance of the power MOSFET. The on-chip power dissipation can be a significant concern when a large power MOSFET is being driven at a high frequency and the VIN voltage is high. It is important to limit the power dissipation through selection of MOSFET and/or operating frequency so the LT3757 does not exceed its maximum junction temperature rating. The junction temperature TJ can be estimated using the following equations: TJ = TA + PIC * JA TA = ambient temperature JA = junction-to-ambient thermal resistance PIC = IC power consumption = VIN * (IQ + IDRIVE) IQ = VIN operation IQ = 1.6mA IDRIVE = average gate drive current = f * QG f = switching frequency QG = power MOSFET total gate charge The LT3757 uses packages with an Exposed Pad for enhanced thermal conduction. With proper soldering to the Exposed Pad on the underside of the package and a full copper plane underneath the device, thermal resistance (JA) will be about 43C/W for the DD package and 40C/W for the MSE package. For an ambient board temperature of TA = 70C and maximum junction temperature of 125C, the maximum IDRIVE (IDRIVE(MAX)) of the DD package can be calculated as: IDRIVE(MAX ) = ( TJ - T A ) 1.28 W - IQ = - 1.6mA VIN ( JA * VIN )
The LT3757 has an internal INTVCC IDRIVE current limit function to protect the IC from excessive on-chip power dissipation. The IDRIVE current limit decreases as the VIN increases (see the INTVCC Minimum Output Current vs VIN graph in the Typical Performance Characteristics section). If IDRIVE reaches the current limit, INTVCC voltage will fall and may trigger the soft-start. Based on the preceding equation and the INTVCC Minimum Output Current vs VIN graph, the user can calculate the maximum MOSFET gate charge the LT3757 can drive at a given VIN and switch frequency. A plot of the maximum QG vs VIN at different frequencies to guarantee a minimum 4.5V INTVCC is shown in Figure 2. As illustrated in Figure 2, a tradeoff between the operating frequency and the size of the power MOSFET may be needed in order to maintain a reliable IC junction temperature. Prior to lowering the operating frequency, however, be sure to check with power MOSFET manufacturers for their most recent low QG, low RDS(ON) devices. Power MOSFET manufacturing technologies are continually improving, with newer and better performance devices being introduced almost yearly.
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APPLICATIONS INFORMATION
300 250 300kHz 200 QG (nC) GND 150 100 1MHz 50 0 0 5 10 15 20 VIN (V) 25 30 35 40
3757 F02 3757 F03
LT3757 INTVCC
DVCC
RVCC
VOUT
CVCC 4.7F
Figure 3. Connecting INTVCC to VOUT
or not the INTVCC pin is connected to an external voltage source, it is always necessary to have the driver circuitry bypassed with a 4.7F low ESR ceramic capacitor to ground immediately adjacent to the INTVCC and GND pins. Operating Frequency and Synchronization The choice of operating frequency may be determined by on-chip power dissipation, otherwise it is a tradeoff between efficiency and component size. Low frequency operation improves efficiency by reducing gate drive current and MOSFET and diode switching losses. However, lower frequency operation requires a physically larger inductor. Switching frequency also has implications for loop compensation. The LT3757 uses a constant-frequency architecture that can be programmed over a 100kHz to 1000kHz range with a single external resistor from the RT pin to ground, as shown in Figure 1. The RT pin must have an external resistor to GND for proper operation of the LT3757. A table for selecting the value of RT for a given operating frequency is shown in Table 1.
Table 1. Timing Resistor (RT ) Value
OSCILLATOR FREQUENCY (kHz) 100 200 300 400 500 600 700 800 900 1000 RT (k) 140 63.4 41.2 30.9 24.3 19.6 16.5 14 12.1 10.5
Figure 2. Recommended Maximum QG vs VIN at Different Frequencies to Ensure INTVCC Higher Than 4.5V
An effective approach to reduce the power consumption of the internal LDO for gate drive is to tie the INTVCC pin to an external voltage source high enough to turn off the internal LDO regulator. If the input voltage VIN does not exceed the absolute maximum rating of both the power MOSFET gate-source voltage (VGS) and the INTVCC overvoltage lockout threshold voltage (17.5V), the INTVCC pin can be shorted directly to the VIN pin. In this condition, the internal LDO will be turned off and the gate driver will be powered directly from the input voltage, VIN. With the INTVCC pin shorted to VIN, however, a small current (around 16A) will load the INTVCC in shutdown mode. For applications that require the lowest shutdown mode input supply current, do not connect the INTVCC pin to VIN. In SEPIC or flyback applications, the INTVCC pin can be connected to the output voltage VOUT through a blocking diode, as shown in Figure 3, if VOUT meets the following conditions: 1. VOUT < VIN (pin voltage) 2. 7.2 < VOUT < 17.5V 3. VOUT < maximum VGS rating of power MOSFET A resistor RVCC can be connected, as shown in Figure 3, to limit the inrush current from VOUT. Regardless of whether
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LT3757
APPLICATIONS INFORMATION
The operating frequency of the LT3757 can be synchronized to an external clock source. By providing a digital clock signal into the SYNC pin, the LT3757 will operate at the SYNC clock frequency. If this feature is used, an RT resistor should be chosen to program a switching frequency 20% slower than SYNC pulse frequency. The SYNC pulse should have a minimum pulse width of 200ns. Tie the SYNC pin to GND if this feature is not used. Duty Cycle Consideration Switching duty cycle is a key variable defining converter operation. As such, its limits must be considered. Minimum on-time is the smallest time duration that the LT3757 is capable of turning on the power MOSFET. This time is generally about 220ns (typical) (see Minimum On-Time in the Electrical Characteristics table). In each switching cycle, the LT3757 keeps the power switch off for at least 220ns (typical) (see Minimum Off-Time in the Electrical Characteristics table). The minimum on-time and minimum off-time and the switching frequency define the minimum and maximum switching duty cycles a converter is able to generate: Minimum duty cycle = minimum on-time * frequency Maximum duty cycle = 1 - (minimum off-time * frequency) Programming the Output Voltage The output voltage (VOUT) is set by a resistor divider, as shown in Figure 1. The positive and negative VOUT are set by the following equations: R2 VOUT, POSITIVE = 1.6 V * 1+ R1 R2 VOUT, NEGATIVE = -0.8 V * 1+ R1 The resistors R1 and R2 are typically chosen so that the error caused by the current flowing into the FBX pin during normal operation is less than 1% (this translates to a maximum value of R1 at about 158k). Soft-Start The LT3757 contains several features to limit peak switch currents and output voltage (VOUT) overshoot during start-up or recovery from a fault condition. The primary purpose of these features is to prevent damage to external components or the load. High peak switch currents during start-up may occur in switching regulators. Since VOUT is far from its final value, the feedback loop is saturated and the regulator tries to charge the output capacitor as quickly as possible, resulting in large peak currents. A large surge current may cause inductor saturation or power switch failure. The LT3757 addresses this mechanism with the SS pin. As shown in Figure 1, the SS pin reduces the power MOSFET current by pulling down the VC pin through Q2. In this way the SS allows the output capacitor to charge gradually toward its final value while limiting the start-up peak currents. The typical start-up waveforms are shown in the Typical Performance Characteristics section. The inductor current IL slewing rate is limited by the soft-start function. Besides start-up, soft-start can also be triggered by the following faults: 1. INTVCC > 17.5V 2. INTVCC < 2.6V 3. Thermal lockout Any of these three faults will cause the LT3757 to stop switching immediately. The SS pin will be discharged by Q3. When all faults are cleared and the SS pin has been discharged below 0.2V, a 10A current source IS2 starts charging the SS pin, initiating a soft-start operation. The soft-start interval is set by the soft-start capacitor selection according to the equation: TSS = CSS * 1.25V 10A
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component values and the operating conditions (including the input voltage, load current, etc.). To compensate the feedback loop of the LT3757, a series resistor-capacitor network is usually connected from the VC pin to GND. Figure 1 shows the typical VC compensation network. For most applications, the capacitor should be in the range of 470pF to 22nF and the resistor should be in the range of , 5k to 50k. A small capacitor is often connected in parallel with the RC compensation network to attenuate the VC voltage ripple induced from the output voltage ripple through the internal error amplifier. The parallel capacitor usually ranges in value from 10pF to 100pF A practical . approach to design the compensation network is to start with one of the circuits in this data sheet that is similar to your application, and tune the compensation network to optimize the performance. Stability should then be checked across all operating conditions, including load current, input voltage and temperature. SENSE Pin Programming For control and protection, the LT3757 measures the power MOSFET current by using a sense resistor (RSENSE) between GND and the MOSFET source. Figure 4 shows a typical waveform of the sense voltage (VSENSE) across the sense resistor. It is important to use Kelvin traces between the SENSE pin and RSENSE, and to place the IC GND as close as possible to the GND terminal of the RSENSE for proper operation.
VSENSE VSENSE = VSENSE(MAX) VSENSE(PEAK) t TS
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APPLICATIONS INFORMATION
FBX Frequency Foldback When VOUT is very low during start-up or a short-circuit fault on the output, the switching regulator must operate at low duty cycles to maintain the power switch current within the current limit range, since the inductor current decay rate is very low during switch off time. The minimum on-time limitation may prevent the switcher from attaining a sufficiently low duty cycle at the programmed switching frequency. So, the switch current will keep increasing through each switch cycle, exceeding the programmed current limit. To prevent the switch peak currents from exceeding the programmed value, the LT3757 contains a frequency foldback function to reduce the switching frequency when the FBX voltage is low (see the Normalized Switching Frequency vs FBX graph in the Typical Performance Characteristics section). The typical frequency foldback waveforms are shown in the Typical Performance Characteristics section. The frequency foldback function prevents IL from exceeding the programmed limits because of the minimum on-time. During frequency foldback, external clock synchronization is disabled to prevent interference with frequency reducing operation. Thermal Lockout If LT3757 die temperature reaches 165C (typical), the part will go into thermal lockout. The power switch will be turned off. A soft-start operation will be triggered. The part will be enabled again when the die temperature has dropped by 5C (nominal). Loop Compensation Loop compensation determines the stability and transient performance. The LT3757 uses current mode control to regulate the output which simplifies loop compensation. The optimum values depend on the converter topology, the
VSENSE(MAX)
DTS
Figure 4. The Sense Voltage During a Switching Cycle
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APPLICATIONS INFORMATION
Due to the current limit function of the SENSE pin, RSENSE should be selected to guarantee that the peak current sense voltage VSENSE(PEAK) during steady state normal operation is lower than the SENSE current limit threshold (see the Electrical Characteristics table). Given a 20% margin, VSENSE(PEAK) is set to be 80mV. Then, the maximum switch ripple current percentage can be calculated using the following equation: = VSENSE 80mV - 0.5 * VSENSE APPLICATION CIRCUITS The LT3757 can be configured as different topologies. The first topology to be analyzed will be the boost converter, followed by the flyback, SEPIC and inverting converters. Boost Converter: Switch Duty Cycle and Frequency The LT3757 can be configured as a boost converter for the applications where the converter output voltage is higher than the input voltage. Remember that boost converters are not short-circuit protected. Under a shorted output condition, the inductor current is limited only by the input supply capability. For applications requiring a step-up converter that is short-circuit protected, please refer to the Applications Information section covering SEPIC converters. The conversion ratio as a function of duty cycle is VOUT 1 = VIN 1- D in continuous conduction mode (CCM). For a boost converter operating in CCM, the duty cycle of the main switch can be calculated based on the output voltage (VOUT) and the input voltage (VIN). The maximum duty cycle (DMAX) occurs when the converter has the minimum input voltage: DMAX =
M1 RFLT SENSE GND CFLT RSENSE
is used in subsequent design examples to calculate inductor value. VSENSE is the ripple voltage across RSENSE. The LT3757 switching controller incorporates 100ns timing interval to blank the ringing on the current sense signal immediately after M1 is turned on. This ringing is caused by the parasitic inductance and capacitance of the PCB trace, the sense resistor, the diode, and the MOSFET. The 100ns timing interval is adequate for most of the LT3757 applications. In the applications that have very large and long ringing on the current sense signal, a small RC filter can be added to filter out the excess ringing. Figure 5 shows the RC filter on SENSE pin. It is usually sufficient to choose 22 for RFLT and 2.2nF to 10nF for CFLT. Keep RFLT's resistance low. Remember that there is 65A flowing out of the SENSE pin. Adding RFLT will affect the SENSE current limit threshold: VSENSE_ILIM = 108mV - 65A * RFLT
GATE LT3757
VOUT - VIN(MIN) VOUT
Discontinuous conduction mode (DCM) provides higher conversion ratios at a given frequency at the cost of reduced efficiencies and higher switching currents.
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Figure 5. The RC Filter on SENSE Pin
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Based on these equations, the user should choose the inductors having sufficient saturation and RMS current ratings. Set the sense voltage at IL(PEAK) to be the minimum of the SENSE current limit threshold with a 20% margin. The sense resistor value can then be calculated to be: RSENSE = 80 mV IL(PEAK )
APPLICATIONS INFORMATION
Boost Converter: Inductor and Sense Resistor Selection For the boost topology, the maximum average inductor current is: 1 IL(MAX ) = IO(MAX ) * 1- DMAX Then, the ripple current can be calculated by: IL = * IL(MAX ) = * IO(MAX ) * 1 1- DMAX
The constant in the preceding equation represents the percentage peak-to-peak ripple current in the inductor, relative to IL(MAX). The inductor ripple current has a direct effect on the choice of the inductor value. Choosing smaller values of IL requires large inductances and reduces the current loop gain (the converter will approach voltage mode). Accepting larger values of IL provides fast transient response and allows the use of low inductances, but results in higher input current ripple and greater core losses. It is recommended that fall within the range of 0.2 to 0.6. Given an operating input voltage range, and having chosen the operating frequency and ripple current in the inductor, the inductor value of the boost converter can be determined using the following equation: L= VIN(MIN) IL * f * DMAX
Boost Converter: Power MOSFET Selection Important parameters for the power MOSFET include the drain-source voltage rating (VDS), the threshold voltage (VGS(TH)), the on-resistance (RDS(ON)), the gate to source and gate to drain charges (QGS and QGD), the maximum drain current (ID(MAX)) and the MOSFET's thermal resistances (RJC and RJA). The power MOSFET will see full output voltage, plus a diode forward voltage, and any additional ringing across its drain-to-source during its off-time. It is recommended to choose a MOSFET whose BVDSS is higher than VOUT by a safety margin (a 10V safety margin is usually sufficient). The power dissipated by the MOSFET in a boost converter is: PFET = I2L(MAX) * RDS(ON) * DMAX + 2 * V2OUT * IL(MAX) * CRSS * f /1A The first term in the preceding equation represents the conduction losses in the device, and the second term, the switching loss. CRSS is the reverse transfer capacitance, which is usually specified in the MOSFET characteristics. For maximum efficiency, RDS(ON) and CRSS should be minimized. From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following equation: TJ = TA + PFET * JA = TA + PFET * (JC + CA)
The peak and RMS inductor current are: IL(PEAK ) = IL(MAX ) * 1+ 2 IL(RMS) = IL(MAX ) * 1+ 2 12
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APPLICATIONS INFORMATION
TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFET temperature in steady state to ensure that absolute maximum ratings are not exceeded. Boost Converter: Output Diode Selection To maximize efficiency, a fast switching diode with low forward drop and low reverse leakage is desirable. The peak reverse voltage that the diode must withstand is equal to the regulator output voltage plus any additional ringing across its anode-to-cathode during the on-time. The average forward current in normal operation is equal to the output current, and the peak current is equal to: ID(PEAK ) = IL(PEAK ) = 1+ * IL(MAX ) 2 It is recommended that the peak repetitive reverse voltage rating VRRM is higher than VOUT by a safety margin (a 10V safety margin is usually sufficient). The power dissipated by the diode is: PD = IO(MAX) * VD and the diode junction temperature is: TJ = TA + PD * RJA The RJA to be used in this equation normally includes the RJC for the device plus the thermal resistance from the board to the ambient temperature in the enclosure. TJ must not exceed the diode maximum junction temperature rating. Boost Converter: Output Capacitor Selection Contributions of ESR (equivalent series resistance), ESL (equivalent series inductance) and the bulk capacitance must be considered when choosing the correct output capacitors for a given output ripple voltage. The effect of these three parameters (ESR, ESL and bulk C) on the output voltage ripple waveform for a typical boost converter is illustrated in Figure 6.
tON tOFF VCOUT VOUT (AC) RINGING DUE TO TOTAL INDUCTANCE (BOARD + CAP)
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VESR
Figure 6. The Output Ripple Waveform of a Boost Converter
The choice of component(s) begins with the maximum acceptable ripple voltage (expressed as a percentage of the output voltage), and how this ripple should be divided between the ESR step VESR and the charging/discharging VCOUT. For the purpose of simplicity, we will choose 2% for the maximum output ripple, to be divided equally between VESR and VCOUT. This percentage ripple will change, depending on the requirements of the application, and the following equations can easily be modified. For a 1% contribution to the total ripple voltage, the ESR of the output capacitor can be determined using the following equation: ESRCOUT 0.01* VOUT ID(PEAK )
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FLYBACK CONVERTER APPLICATIONS The LT3757 can be configured as a flyback converter for the applications where the converters have multiple outputs, high output voltages or isolated outputs. Figure 7 shows a simplified flyback converter. The flyback converter has a very low parts count for multiple outputs, and with prudent selection of turns ratio, can have high output/input voltage conversion ratios with a desirable duty cycle. However, it has low efficiency due to the high peak currents, high peak voltages and consequent power loss. The flyback converter is commonly used for an output power of less than 50W. The flyback converter can be designed to operate either in continuous or discontinuous mode. Compared to continuous mode, discontinuous mode has the advantage of smaller transformer inductances and easy loop compensation, and the disadvantage of higher peak-to-average current and lower efficiency. In the high output voltage applications, the flyback converters can be designed to operate in discontinuous mode to avoid using large transformers.
SUGGESTED RCD SNUBBER VIN NP:NS D
APPLICATIONS INFORMATION
For the bulk C component, which also contributes 1% to the total ripple: COUT IO(MAX ) 0.01* VOUT * f
The output capacitor in a boost regulator experiences high RMS ripple currents, as shown in Figure 6. The RMS ripple current rating of the output capacitor can be determined using the following equation: IRMS(COUT) IO(MAX ) * DMAX 1- DMAX
Multiple capacitors are often paralleled to meet ESR requirements. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the required RMS current rating. Additional ceramic capacitors in parallel are commonly used to reduce the effect of parasitic inductance in the output capacitor, which reduces high frequency switching noise on the converter output. Boost Converter: Input Capacitor Selection The input capacitor of a boost converter is less critical than the output capacitor, due to the fact that the inductor is in series with the input, and the input current waveform is continuous. The input voltage source impedance determines the size of the input capacitor, which is typically in the range of 10F to 100F A low ESR capacitor . is recommended, although it is not as critical as for the output capacitor. The RMS input capacitor ripple current for a boost converter is: IRMS(CIN) = 0.3 * IL
+
CIN
VSN
- +
CSN RSN LP LS ID
+
+
COUT
-
DSN ISW LT3757 GATE SENSE M
+ -
VDS
RSENSE GND
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Figure 7. A Simplified Flyback Converter
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APPLICATIONS INFORMATION
Flyback Converter: Switch Duty Cycle and Turns Ratio The flyback converter conversion ratio in the continuous mode operation is: VOUT NS D = * VIN NP 1- D where NS/NP is the second to primary turns ratio. Figure 8 shows the waveforms of the flyback converter in discontinuous mode operation. During each switching period TS, three subintervals occur: DTS, D2TS, D3TS. During DTS, M is on, and D is reverse-biased. During D2TS, M is off, and LS is conducting current. Both LP and LS currents are zero during D3TS. The flyback converter conversion ratio in the discontinuous mode operation is: VOUT NS D = * VIN NP D2
VDS
According to the preceding equations, the user has relative freedom in selecting the switch duty cycle or turns ratio to suit a given application. The selections of the duty cycle and the turns ratio are somewhat iterative processes, due to the number of variables involved. The user can choose either a duty cycle or a turns ratio as the start point. The following trade-offs should be considered when selecting the switch duty cycle or turns ratio, to optimize the converter performance. A higher duty cycle affects the flyback converter in the following aspects: * Lower MOSFET RMS current ISW(RMS), but higher MOSFET VDS peak voltage * Lower diode peak reverse voltage, but higher diode RMS current ID(RMS) * Higher transformer turns ratio (NP/NS) The choice, D 1 = D + D2 3 (for discontinuous mode operation) gives the power MOSFET the lowest power stress (the product of RMS current and peak voltage). However, in the high output voltage applications, a higher duty cycle may be adopted to limit the large peak reverse voltage of the diode. The choice,
ISW
ISW(MAX)
D 2 = D + D2 3 (for discontinuous mode operation) gives the diode the lowest power stress (the product of RMS current and peak voltage). An extreme high or low duty cycle results in high power stress on the MOSFET or diode, and reduces efficiency. It is recommended to choose a duty cycle between 20% and 80%.
ID
ID(MAX) DTS D2TS TS D3TS t
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Figure 8. Waveforms of the Flyback Converter in Discontinuous Mode Operation
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According to Figure 8, the primary and secondary peak currents are: ILP(PEAK) = ISW(PEAK) = 2 * ILP(MAX) ILS(PEAK) = ID(PEAK) = 2 * ILS(MAX) The primary and second inductor values of the flyback converter transformer can be determined using the following equations: LP = D2MAX * V 2IN(MAX ) * 2 * POUT(MAX ) * f
APPLICATIONS INFORMATION
Flyback Converter: Transformer Design for Discontinuous Mode Operation The transformer design for discontinuous mode of operation is chosen as presented here. According to Figure 8, the minimum D3 (D3MIN) occurs when the converter has the minimum VIN and the maximum output power (POUT). Choose D3MIN to be equal to or higher than 10% to guarantee the converter is always in discontinuous mode operation. Then the maximum duty cycle (DMAX) can be calculated using the following equation: DMAX = (1- D3MIN ) * VOUT * NP VIN * NS + VOUT * NP
The maximum average primary currents can be calculated by the following equation: ILP(MAX ) = ISW(MAX ) = POUT(MAX ) DMAX * VIN(MIN) *
D22 *( VOUT + VD) LS = 2 * IOUT(MAX ) * f The primary to second turns ratio is: NSP L SP = NSN L SN Flyback Converter: Snubber Design Transformer leakage inductance (on either the primary or secondary) causes a voltage spike to occur after the MOSFET turn-off. This is increasingly prominent at higher load currents, where more stored energy must be dissipated. In some cases a snubber circuit will be required to avoid overvoltage breakdown at the MOSFET's drain node. There are different snubber circuits, and Application Note 19 is a good reference on snubber design. An RCD snubber is shown in Figure 7. The snubber resistor value (RSN) can be calculated by the following equation: V 2SN - VSN * VOUT * RSN = 2 * NSP NSN
where is the converter efficiency. If the flyback converter has multiple outputs, POUT(MAX) is the sum of all the output power. The maximum average secondary current is: ILS(MAX ) = ID(MAX ) = where: D2 = 1 - DMAX - D3 the primary and secondary RMS currents are: ILP(RMS) = 2 * ILP(MAX ) * ILS(RMS) = 2 * ILS(MAX ) * DMAX 3 D2 3 IOUT(MAX ) D2
I2SW(PEAK ) * L LK * f
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APPLICATIONS INFORMATION
where VSN is the snubber capacitor voltage. A smaller VSN results in a larger snubber loss. A reasonable VSN is 2 to 2.5 times of: VOUT * NSP NSN LLK is the leakage inductance of the primary winding, which is usually specified in the transformer characteristics. LLK can be obtained by measuring the primary inductance with the secondary windings shorted. The snubber capacitor value (CCN) can be determined using the following equation: CCN = VSN VSN * RCN * f Flyback Converter: Power MOSFET Selection For the flyback configuration, the MOSFET is selected with a VDC rating high enough to handle the maximum VIN, the reflected secondary voltage and the voltage spike due to the leakage inductance. Approximate the required MOSFET VDC rating using: BVDSS > k * VDS(PEAK) where:
VDS(PEAK ) = VIN(MAX ) + VSN
A reasonable factor for k is 1.3, which provides an overall 30% margin. The power dissipated by the MOSFET in a flyback converter is: PFET = I2M(RMS) * RDS(ON) + 2 * V2DS(PEAK) * IL(MAX) * CRSS * f /1A The first term in this equation represents the conduction losses in the device, and the second term, the switching loss. CRSS is the reverse transfer capacitance, which is usually specified in the MOSFET characteristics. From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following equation: TJ = TA + PFET * JA = TA + PFET * (JC + CA) TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFET temperature in steady state to ensure that absolute maximum ratings are not exceeded.
where VSN is the voltage ripple across CCN. A reasonable VSN is 5% to 10% of VSN. The reverse voltage rating of DSN should be higher than the sum of VSN and VIN(MAX). Flyback Converter: Sense Resistor Selection In a flyback converter, when the power switch is turned on, the current flowing through the sense resistor (ISENSE) is: ISENSE = ILP Set the sense voltage at ILP(PEAK) to be the minimum of the SENSE current limit threshold with a 20% margin. The sense resistor value can then be calculated to be: RSENSE = 80 mV ILP(PEAK )
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Flyback Converter: Input Capacitor Selection The input capacitor in a flyback converter is subject to a large RMS current due to the discontinuous primary current. To prevent large voltage transients, use a low ESR input capacitor sized for the maximum RMS current. The RMS ripple current rating of the input capacitors in discontinuous operation can be determined using the following equation:
IRMS(CIN),DISCONTINUOUS POUT(MAX ) VIN(MIN) * * 4 - (3 * DMAX ) 3 * DMAX
APPLICATIONS INFORMATION
Flyback Converter: Output Diode Selection The output diode in a flyback converter is subject to large RMS current and peak reverse voltage stresses. A fast switching diode with a low forward drop and a low reverse leakage is desired. Schottky diodes are recommended if the output voltage is below 100V. Approximate the required peak repetitive reverse voltage rating VRRM using: N VRRM > SN * VIN(MAX ) + VOUT NSP The power dissipated by the diode is: PD = IO(MAX) * VD and the diode junction temperature is: TJ = TA + PD * RJA The RJA to be used in this equation normally includes the RJC for the device, plus the thermal resistance from the board to the ambient temperature in the enclosure. TJ must not exceed the diode maximum junction temperature rating. Flyback Converter: Output Capacitor Selection The output capacitor of the flyback converter has a similar operation condition as that of the boost converter. Refer to the Boost Converter: Output Capacitor Selection section for the calculation of COUT and ESRCOUT. The RMS ripple current rating of the output capacitors in discontinuous operation can be determined using the following equation:
IRMS(COUT ),DISCONTINUOUS IO(MAX ) * 4 - (3 * D2) 3 * D2
SEPIC CONVERTER APPLICATIONS The LT3757 can be configured as a SEPIC (single-ended primary inductance converter), as shown in Figure 1. This topology allows for the input to be higher, equal, or lower than the desired output voltage. The conversion ratio as a function of duty cycle is: VOUT + VD D = VIN 1- D in continuous conduction mode (CCM). In a SEPIC converter, no DC path exists between the input and output. This is an advantage over the boost converter for applications requiring the output to be disconnected from the input source when the circuit is in shutdown. Compared to the flyback converter, the SEPIC converter has the advantage that both the power MOSFET and the output diode voltages are clamped by the capacitors (CIN, CDC and COUT), therefore, there is less voltage ringing across the power MOSFET and the output diodes. The SEPIC converter requires much smaller input capacitors than those of the flyback converter. This is due to the fact
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APPLICATIONS INFORMATION
that, in the SEPIC converter, the inductor L1 is in series with the input, and the ripple current flowing through the input capacitor is continuous. SEPIC Converter: Switch Duty Cycle and Frequency For a SEPIC converter operating in CCM, the duty cycle of the main switch can be calculated based on the output voltage (VOUT), the input voltage (VIN) and the diode forward voltage (VD). The maximum duty cycle (DMAX) occurs when the converter has the minimum input voltage: DMAX = VOUT + VD VIN(MIN) + VOUT + VD In a SEPIC converter, the switch current is equal to IL1 + IL2 when the power switch is on, therefore, the maximum average switch current is defined as: ISW(MAX ) = IL1(MAX ) + IL 2(MAX ) = IO(MAX ) * and the peak switch current is: 1 ISW(PEAK ) = 1+ * IO(MAX ) * 1- DMAX 2 The constant in the preceding equations represents the percentage peak-to-peak ripple current in the switch, relative to ISW(MAX), as shown in Figure 9. Then, the switch ripple current ISW can be calculated by: ISW = * ISW(MAX) The inductor ripple currents IL1 and IL2 are identical: IL1 = IL2 = 0.5 * ISW The inductor ripple current has a direct effect on the choice of the inductor value. Choosing smaller values of IL requires large inductances and reduces the current loop gain (the converter will approach voltage mode). Accepting larger values of IL allows the use of low inductances, but results in higher input current ripple and greater core losses. It is recommended that falls in the range of 0.2 to 0.4. 1 1- DMAX
SEPIC Converter: Inductor and Sense Resistor Selection As shown in Figure 1, the SEPIC converter contains two inductors: L1 and L2. L1 and L2 can be independent, but can also be wound on the same core, since identical voltages are applied to L1 and L2 throughout the switching cycle. For the SEPIC topology, the current through L1 is the converter input current. Based on the fact that, ideally, the output power is equal to the input power, the maximum average inductor currents of L1 and L2 are: IL1(MAX ) = IIN(MAX ) = IO(MAX ) * IL2(MAX ) =IO(MAX )
ISW ISW =
DMAX 1- DMAX
ISW(MAX)
ISW(MAX)
DTS TS
t
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Figure 9. The Switch Current Waveform of the SEPIC Converter
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Based on the preceding equations, the user should choose the inductors having sufficient saturation and RMS current ratings. In a SEPIC converter, when the power switch is turned on, the current flowing through the sense resistor (ISENSE) is the switch current. Set the sense voltage at ISENSE(PEAK) to be the minimum of the SENSE current limit threshold with a 20% margin. The sense resistor value can then be calculated to be: RSENSE = 80 mV ISW(PEAK )
APPLICATIONS INFORMATION
Given an operating input voltage range, and having chosen the operating frequency and ripple current in the inductor, the inductor value (L1 and L2 are independent) of the SEPIC converter can be determined using the following equation: L1= L2 = VIN(MIN) 0.5 * ISW * f * DMAX
For most SEPIC applications, the equal inductor values will fall in the range of 1H to 100H. By making L1 = L2, and winding them on the same core, the value of inductance in the preceding equation is replaced by 2L, due to mutual inductance: L= VIN(MIN) ISW * f * DMAX
SEPIC Converter: Power MOSFET Selection For the SEPIC configuration, choose a MOSFET with a VDC rating higher than the sum of the output voltage and input voltage by a safety margin (a 10V safety margin is usually sufficient). The power dissipated by the MOSFET in a SEPIC converter is: PFET = I2SW(MAX) * RDS(ON) * DMAX + 2 * (VIN(MIN) + VOUT)2 * IL(MAX) * CRSS * f /1A The first term in this equation represents the conduction losses in the device, and the second term, the switching loss. CRSS is the reverse transfer capacitance, which is usually specified in the MOSFET characteristics. For maximum efficiency, RDS(ON) and CRSS should be minimized. From a known power dissipated in the power MOSFET, its junction temperature can be obtained using the following equation: TJ = TA + PFET * JA = TA + PFET * (JC + CA) TJ must not exceed the MOSFET maximum junction temperature rating. It is recommended to measure the MOSFET temperature in steady state to ensure that absolute maximum ratings are not exceeded.
This maintains the same ripple current and energy storage in the inductors. The peak inductor currents are: IL1(PEAK) = IL1(MAX) + 0.5 * IL1 IL2(PEAK) = IL2(MAX) + 0.5 * IL2 The RMS inductor currents are: 2 IL1(RMS) = IL1(MAX ) * 1+ L1 12 where: L 1 = IL1 IL1(MAX )
2L2 IL2(RMS) = IL2(MAX ) * 1+ 12 where: L 2 = IL2 IL2 (MAX )
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APPLICATIONS INFORMATION
SEPIC Converter: Output Diode Selection To maximize efficiency, a fast switching diode with a low forward drop and low reverse leakage is desirable. The average forward current in normal operation is equal to the output current, and the peak current is equal to: 1 ID(PEAK ) = 1+ * IO(MAX ) * 1- DMAX 2 It is recommended that the peak repetitive reverse voltage rating VRRM is higher than VOUT + VIN(MAX) by a safety margin (a 10V safety margin is usually sufficient). The power dissipated by the diode is: PD = IO(MAX) * VD and the diode junction temperature is: TJ = TA + PD * RJA The RJA used in this equation normally includes the RJC for the device, plus the thermal resistance from the board, to the ambient temperature in the enclosure. TJ must not exceed the diode maximum junction temperature rating. SEPIC Converter: Output and Input Capacitor Selection The selections of the output and input capacitors of the SEPIC converter are similar to those of the boost converter. Please refer to the Boost Converter, Output Capacitor Selection and Boost Converter, Input Capacitor Selection sections. SEPIC Converter: Selecting the DC Coupling Capacitor The DC voltage rating of the DC coupling capacitor (CDC, as shown in Figure 1) should be larger than the maximum input voltage: VCDC > VIN(MAX) CDC has nearly a rectangular current waveform. During the switch off-time, the current through CDC is IIN, while approximately -IO flows during the on-time. The RMS rating of the coupling capacitor is determined by the following equation: IRMS(CDC) > IO(MAX ) * VOUT + VD VIN(MIN)
A low ESR and ESL, X5R or X7R ceramic capacitor works well for CDC. INVERTING CONVERTER APPLICATIONS The LT3757 can be configured as a dual-inductor inverting topology, as shown in Figure 10. The VOUT to VIN ratio is: VOUT - VD D =- VIN 1- D in continuous conduction mode (CCM).
L1 VIN
+
CDC
-
L2
+
CIN LT3757 GATE SENSE RSENSE GND M1 D1 COUT
- +
VOUT
+
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Figure 10. A Simplified Inverting Converter
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After specifying the maximum output ripple, the user can select the output capacitors according to the preceding equation. The ESR can be minimized by using high quality X5R or X7R dielectric ceramic capacitors. In many applications, ceramic capacitors are sufficient to limit the output voltage ripple. The RMS ripple current rating of the output capacitor needs to be greater than: IRMS(COUT) > 0.3 * IL2 Inverting Converter: Selecting the DC Coupling Capacitor The DC voltage rating of the DC coupling capacitor (CDC, as shown in Figure 10) should be larger than the maximum input voltage minus the output voltage (negative voltage): VCDC > VIN(MAX) - VOUT CDC has nearly a rectangular current waveform. During the switch off-time, the current through CDC is IIN, while approximately -IO flows during the on-time. The RMS rating of the coupling capacitor is determined by the following equation: DMAX IRMS(CDC) > IO(MAX ) * 1- DMAX A low ESR and ESL, X5R or X7R ceramic capacitor works well for CDC.
APPLICATIONS INFORMATION
Inverting Converter: Switch Duty Cycle and Frequency For an inverting converter operating in CCM, the duty cycle of the main switch can be calculated based on the negative output voltage (VOUT) and the input voltage (VIN). The maximum duty cycle (DMAX) occurs when the converter has the minimum input voltage: DMAX = VOUT - VD VOUT - VD - VIN(MIN)
Inverting Converter: Inductor, Sense Resistor, Power MOSFET, Output Diode and Input Capacitor Selections The selections of the inductor, sense resistor, power MOSFET, output diode and input capacitor of an inverting converter are similar to those of the SEPIC converter. Please refer to the corresponding SEPIC converter sections. Inverting Converter: Output Capacitor Selection The inverting converter requires much smaller output capacitors than those of the boost, flyback and SEPIC converters for similar output ripples. This is due to the fact that, in the inverting converter, the inductor L2 is in series with the output, and the ripple current flowing through the output capacitors are continuous. The output ripple voltage is produced by the ripple current of L2 flowing through the ESR and bulk capacitance of the output capacitor: 1 VOUT(P - P) = IL2 * ESRCOUT + 8 * f * COUT
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APPLICATIONS INFORMATION
Board Layout The high speed operation of the LT3757 demands careful attention to board layout and component placement. The Exposed Pad of the package is the only GND terminal of the IC, and is important for thermal management of the IC. Therefore, it is crucial to achieve a good electrical and thermal contact between the Exposed Pad and the ground plane of the board. For the LT3757 to deliver its full output power, it is imperative that a good thermal path be provided to dissipate the heat generated within the package. It is recommended that multiple vias in the printed circuit board be used to conduct heat away from the IC and into a copper plane with as much area as possible. To prevent radiation and high frequency resonance problems, proper layout of the components connected to the IC is essential, especially the power paths with higher di/dt. The following high di/dt loops of different topologies should be kept as tight as possible to reduce inductive ringing: * In boost configuration, the high di/dt loop contains the output capacitor, the sensing resistor, the power MOSFET and the Schottky diode. * In flyback configuration, the high di/dt primary loop contains the input capacitor, the primary winding, the power MOSFET and the sensing resistor. The high di/dt secondary loop contains the output capacitor, the secondary winding and the output diode. * In SEPIC configuration, the high di/dt loop contains the power MOSFET, sense resistor, output capacitor, Schottky diode and the coupling capacitor. * In inverting configuration, the high di/dt loop contains power MOSFET, sense resistor, Schottky diode and the coupling capacitor.
CIN
VIN
CC1 CC2 R1 R2 RSS RT RC 1 LT3757 2 3 4 5 10
9
R3 R4
L1
8 7 6
CVCC
1 2 M1 3 4
8 7 6 5
RS VIAS TO GROUND PLANE
COUT2
COUT1
D1
VOUT
3757 F10
Figure 11. 8V to 16V Input, 24V/2A Output Boost Converter Suggested Layout
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LT3757
sensing resistor divider should connect independently to the top of the output capacitor (Kelvin connection), staying away from any high dV/dt traces. Place the divider resistors near the LT3757 in order to keep the high impedance FBX node short. Figure 11 shows the suggested layout of the 8V to 16V Input, 24V/2A Output Boost Converter. Recommended Component Manufacturers Some of the recommended component manufacturers are listed in Table 2.
APPLICATIONS INFORMATION
Check the stress on the power MOSFET by measuring its drain-to-source voltage directly across the device terminals (reference the ground of a single scope probe directly to the source pad on the PC board). Beware of inductive ringing, which can exceed the maximum specified voltage rating of the MOSFET. If this ringing cannot be avoided, and exceeds the maximum rating of the device, either choose a higher voltage device or specify an avalancherated power MOSFET. The small-signal components should be placed away from high frequency switching nodes. For optimum load regulation and true remote sensing, the top of the output voltage
Table 2. Recommended Component Manufacturers
VENDOR AVX BH Electronics Coilcraft Coiltronics Diodes, Inc Fairchild General Semiconductor International Rectifier IRC Kemet Magnetics Inc Microsemi Murata-Erie Nichicon On Semiconductor Panasonic Sanyo Sumida Taiyo Yuden TDK Thermalloy Tokin Toko United Chemicon Vishay/Dale Vishay/Siliconix Vishay/Sprague Wurth Electronik Zetex COMPONENTS Capacitors Inductors, Transformers Inductors Inductors Diodes MOSFETs Diodes MOSFETs, Diodes Sense Resistors Capacitors Toroid Cores Diodes Inductors, Capacitors Capacitors Diodes Capacitors Capacitors Inductors Capacitors Capacitors, Inductors Heat Sinks Capacitors Inductors Capacitors Resistors MOSFETs Capacitors Inductors Small-Signal Discretes TELEPHONE (207) 282-5111 (952) 894-9590 (847) 639-6400 (407) 241-7876 (805) 446-4800 (408) 822-2126 (516) 847-3000 (310) 322-3331 (361) 992-7900 (408) 986-0424 (800) 245-3984 (617) 926-0404 (770) 436-1300 (847) 843-7500 (602) 244-6600 (714) 373-7334 (619) 661-6835 (847) 956-0667 (408) 573-4150 (562) 596-1212 (972) 243-4321 (408) 432-8020 (847) 699-3430 (847) 696-2000 (605) 665-9301 (800) 554-5565 (207) 324-4140 (605) 886-4385 (631) 543-7100 WEB ADDRESS avx.com bhelectronics.com coilcraft.com coiltronics.com diodes.com fairchildsemi.com generalsemiconductor.com irf.com irctt.com kemet.com mag-inc.com microsemi.com murata.co.jp nichicon.com onsemi.com panasonic.com sanyo.co.jp sumida.com t-yuden.com component.tdk.com aavidthermalloy.com nec-tokinamerica.com tokoam.com chemi-com.com vishay.com vishay.com vishay.com we-online.com zetex.com
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TYPICAL APPLICATIONS
3.3V Input, 5V/10A Output Boost Converter
L1 0.5H CIN 22F 6.3V 2 VIN INTVCC SHDN/UVLO 34k SYNC
VIN 3.3V
49.9k
LT3757
GATE
CVCC 4.7F 10V X5R
D1
VOUT 5V 10A 34k 1%
M1
FBX RT SS 41.2k 300kHz 0.1F VC SENSE GND
22
+
COUT1 150F 6.3V 4 COUT2 22F 6.3V X5R 4
6.8k 22nF 2.2nF
0.004 1W 15.8k 1%
CIN: TAIYO YUDEN JMK325BJ226MM COUT1: PANASONIC EEFUEOJ151R COUT2: TAIYO YUDEN JMK325BJ226MM
D1: MBRB2515L L1: VISHAY SILICONIX IHLP-5050FD-01 M1: VISHAY SILICONIX SI4448DY
3757 TA02a
Efficiency vs Output Current
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 0.001 0.01 0.1 1 10
3757 TA02b
OUTPUT CURRENT (A)
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8V to 16V Input, 24V/2A Output Boost Converter
VIN 8V TO 16V
TYPICAL APPLICATIONS
CIN 10F 25V X5R
R3 200k R4 43.2k SYNC
VIN SHDN/UVLO
L1 10H D1 VOUT 24V 2A R2 226k 1%
LT3757
GATE SENSE RT SS VC GND RC 22k CC1 6.8nF M1
+
FBX INTVCC RS 0.01 1W
COUT1 47F 35V 4
RT 41.2k 300kHz
CC2 100pF CSS 0.1F
CVCC 4.7F 10V X5R
R1 16.2k 1%
COUT2 10F 25V X5R
CIN, COUT2: MURATA GRM31CR61E106KA12 COUT1: KEMET T495X476K035AS D1: ON SEMI MBRS340T3G L1: VISHAY SILICONIX IHLP-5050FD-01 10H M1: VISHAY SILICONIX Si4840BDP
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Efficiency vs Output Current
100 90 80 EFFICIENCY (%) 70 60 50 40 30 0.001 IOUT 1.6A 1A/DIV 0.4A VIN = 8V VIN = 16V VOUT 500mV/DIV (AC)
Load Step Response at VIN = 12V
500s/DIV 0.1 1 0.01 OUTPUT CURRENT (A) 10
3757 TA03b
3757 TA03c
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LT3757
TYPICAL APPLICATIONS
High Voltage Flyback Power Supply
DANGER! High Voltage Operation by High Voltage Trained Personnel Only VIN 5V TO 12V T1 1:10 CIN 150F 6.3V 2 105k SHDN/UVLO 46.4k SYNC VIN INTVCC CVCC 47F 25V X5R D1 VOUT 350V 10mA 1.50M 1% 1M 1% VSW M1 22 GND SENSE FBX 10nF 6.8k 0.1F 100pF
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* *
0.02 VOUT 2V/DIV (AC) VSW 20V/DIV
LT3757
GATE
1M 1%
COUT 68nF 2
RT SS VC 140k 100kHz
16.2k 1%
22nF
CIN : MURATA GRM32DR61C106K COUT : TDK C3225X7R2J683K D1: VISHAY SILICONIX GSD2004S DUAL DIODE CONNECTED IN SERIES M1: VISHAY SILICONIX Si7850DP T1: TDK DCT15EFD-U44S003
Start-Up Waveforms
Switching Waveforms
VOUT 100V/DIV
2ms/DIV
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5s/DIV
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LT3757
5.5V to 36V Input, 12V/2A Output SEPIC Converter
TYPICAL APPLICATIONS
VIN 5.5V TO 36V CIN 4.7F 50V 2
105k
*
VIN SHDN/UVLO L1A
IL1A
CDC 4.7F 50V, X5R, 2
D1
46.4k SYNC
LT3757
GATE SENSE RT SS VC
VSW M1 IL1B L1B
VOUT 12V 2A 105k 1%
*
0.008 1W
+
FBX GND INTVCC
COUT1 47F 20V 2
41.2k 300kHz 0.1F
10k 6.8nF 4.7F 10V X5R 15.8k 1%
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COUT2 10F 25V X5R
CIN, CDC: TAIYO YUDEN UMK316BJ475KL COUT1: KEMET T495X476K020AS COUT2: TAIYO YUDEN TMK432BJ106MM D1: ON SEMI MBRS360T3G L1A, L1B: COILTRONICS DRQ127-3R3 (*COUPLED INDUCTORS) M1: VISHAY SILICONIX Si7460DP
Efficiency vs Output Current
100 90 VIN = 8V 80 EFFICIENCY (%) 70 60 50 40 30 20 0.001 0.1 1 0.01 OUTPUT CURRENT (A) 10
3757 TA05b
Load Step Waveforms
VIN = 16V
VOUT 200mV/DIV (AC)
IOUT 1.6A 1A/DIV 0.4A 500s/DIV
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Start-Up Waveforms
VIN = 12V
Frequency Foldback Waveforms When Output Short-Circuits
VOUT 10V/DIV VIN = 12V
VOUT 5V/DIV
VSW 20V/DIV
IL1A + IL1B 5A/DIV 2ms/DIV
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IL1A + IL1B 5A/DIV
50s/DIV
3757 TA05e
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LT3757
TYPICAL APPLICATIONS
5V to 12V Input, 12V/0.2A Output SEPIC Converter
VIN 5V TO 12V + CIN2 47F 16V CIN1 1F 16V, X5R 105k VIN SHDN/UVLO 46.4k SYNC LT3757 GATE SENSE RT SS VC 30.9k 400kHz 0.1F 100pF GND INTVCC FBX 0.02 D2 6 * * T1 1,2,3,4 CDC1 4.7F 16V, X5R
D1 CDC2 4.7F 16V X5R 1.05k 1% 5 * 158 1% COUT2 4.7F 16V, X5R 3 COUT2 4.7F 16V, X5R 3
M1
VOUT1 12V 0.4A
GND
22k 6.8nF
VOUT2 -12V 0.4A
CVCC 4.7F 10V X5R
3757 TA06
D1, D2: MBRS140T3 T1: COILTRONICS VP1-0076 (*PRIMARY = 4 WINDINGS IN PARALLEL) M1: SILICONIX/VISHAY Si4840BDY
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LT3757
Nonisolated Inverting SLIC Supply
VP5-0155 (PRIMARY = 3 WINDINGS IN PARALLEL) D1 DFLS160
TYPICAL APPLICATIONS
VIN 5V TO 16V
CIN 22F 25V, X5R 2
GND R2 105k R1 46.4k SYNC VIN SHDN/UVLO LT3757 GATE SENSE RT SS VC GND INTVCC 0.012 0.5W CVCC 4.7F 10V, X5R 6 * 15.8k 464k
3757 TA07
C2 10F 50V X5R
T1 1,2,3
* 4 * D2 DFLS160
C3 22F 25V X5R
M1 Si7850DP
FBX *
5
C4 22F 25V X5R
VOUT1 -24V 200mA COUT 3.3F 100V
63.4k 200kHz 0.1F 100pF
9.1k 10nF
D3 DFLS160 C5 22F 25V X5R
VOUT1 -72V 200mA
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PACKAGE DESCRIPTION
DD Package 10-Lead Plastic DFN (3mm x 3mm)
(Reference LTC DWG # 05-08-1699)
R = 0.115 TYP 6 0.675 0.05 0.38 0.10 10
3.50 0.05 1.65 0.05 2.15 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 0.05 0.50 BSC 2.38 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE PIN 1 TOP MARK (SEE NOTE 6)
3.00 0.10 (4 SIDES)
1.65 0.10 (2 SIDES)
(DD) DFN 1103
5 0.200 REF 0.75 0.05 2.38 0.10 (2 SIDES)
1 0.25 0.05 0.50 BSC
0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD
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MSE Package 10-Lead Plastic MSOP Exposed Die Pad ,
(Reference LTC DWG # 05-08-1664 Rev C)
BOTTOM VIEW OF EXPOSED PAD OPTION
PACKAGE DESCRIPTION
2.794 (.110
0.102 .004)
0.889 (.035
0.127 .005)
1
2.06 0.102 (.081 .004) 1.83 0.102 (.072 .004)
0.29 REF
5.23 (.206) MIN
2.083 (.082
0.102 3.20 - 3.45 .004) (.126 - .136)
0.05 REF DETAIL "B" CORNER TAIL IS PART OF DETAIL "B" THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE 0.497 0.076 (.0196 .003) REF
10
0.50 0.305 0.038 (.0197) (.0120 .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT
3.00 0.102 (.118 .004) (NOTE 3)
10 9 8 7 6
4.90 0.152 (.193 .006) 0.254 (.010) GAUGE PLANE 0.53 0.152 (.021 .006) DETAIL "A" 0.18 (.007) SEATING PLANE 1.10 (.043) MAX DETAIL "A" 0 - 6 TYP 12345
3.00 0.102 (.118 .004) (NOTE 4)
0.86 (.034) REF
0.17 -0.27 (.007 - .011) TYP
NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.50 (.0197) BSC
0.1016 (.004
0.0508 .002)
MSOP (MSE) 0908 REV C
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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LT3757
TYPICAL APPLICATION
High Efficiency Inverting Power Supply
VIN 5V TO 15V 100 CIN 47F 16V X5R R2 105k R1 46.4k SYNC VIN SHDN/UVLO
Efficiency vs Output Current
90 L2 VOUT -5V 3A to 5A 80 EFFICIENCY (%) 70 60 50 40 30 16k COUT 100F 6.3V, X5R 2
3757 TA08a
*
L1
CDC 47F 25V, X5R
LT3757
GATE SENSE RT SS VC GND 9.1k 10nF FBX INTVCC CVCC 4.7F 10V X5R M1 Si7848BDP 0.006 1W 84.5k D1 MBRD835L
*
VIN = 5V VIN = 16V
41.2k 300kHz 0.1F
20 10 0.001 0.1 1 0.01 OUTPUT CURRENT (A) 10
3757 TA08b
L1, L2: COILTRONICS DRQ127-3R3 (*COUPLED INDUCTORS)
RELATED PARTS
PART NUMBER LT1175 LT1619 LTC1624 LTC1700 LTC1871, LTC1871-1 LTC1871-7 LT1930 LT1931 LTC3401/LTC3402 LTC3704 LT3724 LT3782A DESCRIPTION Negative Linear Low Dropout Regulator Current Mode PWM Controller Current Mode DC/DC Controller No RSENSE Synchronous Step-Up Controller No RSENSETM Boost, Flyback and SEPIC Controller Wide Input Range Controller 1.2MHz, SOT-23 Boost Converter Inverting 1.2MHz, SOT-23 Converter 1A/2A 3MHz Synchronous Boost Converters Positive-to-Negative DC/DC Controller High Voltage, Current Mode Switching Regulator Controller 2-Phase Step-Up DC/DC Controller COMMENTS User-Selectable Current Limit from 200mA to 800mA, 0.4V Dropout at 500mA, 45A Operating Current 300kHz Fixed Frequency, Boost, SEPIC, Flyback Topology; 1.9V VIN 18V VIN Up to 36V, 300kHz Operating Frequency; Buck, Boost, SEPIC Design, SO-8 Package Up to 95% Efficiency, Operation as Low as 0.9V Input 2.5V VIN 30V, Current Mode Control, Programmable fOSC from 50kHz to 1MHz No RSENSE, 7V Gate Drive, Current Mode Control 2.6V VIN 16V, Up to 34V Output, Miniature Design Positive-to-Negative DC/DC Conversion, Miniature Design 0.5V VIN 5V, Up to 97% Efficiency, Very Small Solution No RSENSE, Current Mode Control, 50kHz to 1MHz Wide Input Range: 4V to 60V 6V VIN 40V, 4A Gate Drive, 150kHz to 500kHz Wide Input Range; Boost, Flyback and SEPIC Topologies VIN and VOUT Limited Only by External Components, Used in Boost, Flyback and SEPIC Topologies Delivers Up to 5A, 550kHz Fixed Frequency, Current Mode 1.23V VOUT 36V, 4V VIN 60V, 120A IQ 4V VIN 36V, Up to 12 Phases Can Be Paralleled and Clocked Out-of-Phase 2.75V VIN 9.8V, Current Mode Control, SOT-23 or 2mm x 3mm DFN-8
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LTC3803, LTC3803-3, Constant-Frequency, Current Mode DC/DC LTC3803-5 Controller in ThinSOTTM LTC3805, LTC3805-5 LTC3822 LT3845 LTC3862/LTC3862-1 LTC3872 Adjustable, Synchronizable Frequency, Current Mode DC/DC Controller No RSENSE SOT-23 Boost Controller Low IQ, High Voltage Single Output Synchronous Step-Down DC/DC Controller Multiphase High Power Step-Up DC/DC Controller No RSENSE Boost, Flyback and SEPIC Controller
No RSENSE and ThinSOT are trademarks of Linear Technology Corporation.
36 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
LT 0509 * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2008

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